Rotating harmonic rejection mixer

ABSTRACT

In one embodiment, the present invention includes a mixer circuit to receive and generate a mixed signal from a radio frequency (RF) signal and a master clock signal, a switch stage coupled to an output of the mixer circuit to rotatingly switch the mixed signal to multiple gain stages coupled to the switch stage, and a combiner to combine an output of the gain stages.

This application is a continuation-in-part of U.S. patent application Ser. No. 12/794,113, filed on Jun. 4, 2010, which in turn is a divisional of U.S. patent application Ser. No. 11/824,417, filed on Jun. 29, 2007, now U.S. Pat. No. 7,756,504, issued Jul. 13, 2010, entitled “A Rotating Harmonic Rejection Mixer,” the contents of which are hereby incorporated by reference.

FIELD OF THE INVENTION

Embodiments of the present invention relate to a mixer and more particularly to a mixer to perform harmonic rejection.

BACKGROUND

A conventional receiver may include at least one mixer to downconvert the frequency of an incoming signal. More specifically, the mixer typically multiplies the incoming wireless signal with a local oscillator signal to produce a signal that has spectral energy that is distributed at sums and differences of the local oscillator and incoming signal's frequencies. For a downconversion mixer, the desired output is the difference between the local oscillator and incoming signal frequency. If the local oscillator signal is a pure sinusoid only the spectral energy of the incoming signal that is at an intermediate frequency (IF) away from the local oscillator (LO) signal appears at the output of the downconversion mixer. However, for certain mixing applications, the local oscillator signal may be a non-sinusoidal, such as a square wave signal, which contains spectral energy that is located at a fundamental frequency and additional spectral energy that is located at harmonic frequencies of the fundamental frequency. Mixing the incoming signal with such a local oscillator signal causes the spectral energy of the incoming signal at IF away from the harmonics of the LO signal to also appear along with the desired signal at the downconversion mixer's output.

Harmonic rejection mixers exist that include multiple mixers such as Gilbert cell type mixers to each receive a scaled version of an incoming signal, where the outputs of each mixer stage are summed to provide a downconverted (or upconverted) output. Each mixer may operate at a phase difference from the other mixers, and each scaling factor that scales the incoming signal may be in accordance with a predetermined sinusoidal function such that the harmonic rejection mixer ideally rejects all harmonics except M×N+/−1, where M is any integer and N is the number of individual mixer stages. However, actual implementations do not operate according to this ideal. Instead, in practical implementations in a semiconductor integrated circuit (IC) process, various problems exist. These problems include difficulties in device matching among the different mixers, as random device mismatches between active devices, i.e., transistors in the mixers may cause the scaling factors to deviate from an ideal value, causing degradation in harmonic rejection. Furthermore, the phases of a LO signal provided to each branch may also deviate, causing harmonic rejection degradation.

To overcome such problems in conventional harmonic rejection mixers, very large device sizes are needed, which creates circuits that are very large and consume significant power. Furthermore, even if a large size is implemented such that the standard deviation of random mismatches is reduced (in turn raising power and area by a factor of 4), harmonic rejection can still be affected by the duty cycle of each LO waveform. Accordingly, positive and negative LO signals should be exactly 180 degrees out of phase, requiring additional well-matched components and operation at higher frequencies, again causing more consumption of power to achieve a desired performance level. In many designs, the amount of harmonic rejection that can realistically be achieved in such a mixer may be between approximately 30-40 dB, when operating at an LO frequency of several hundred MHz. Such performance may be acceptable for some applications. However, operation at this level can cause stricter tolerances for other components in a total budget for a given receiver design.

Thus, there exists a continuing need for a mixer that rejects harmonic frequencies that may be introduced by a local oscillator signal that is not a pure sinusoid.

SUMMARY OF THE INVENTION

According to one aspect, a rotating harmonic rejection mixer can be implemented using a multiple stage design. In one embodiment, the mixer has a first stage including master radio frequency (RF) devices each to receive a common incoming RF signal and provide an RF current, and master local oscillator (LO) devices each coupled to an output of one of the master RF devices, each to receive the RF current and mix the RF current with a master clock signal to obtain a mixed signal. In addition, rotating switch devices are each coupled to one of the master LO devices to cyclically switch the corresponding mixed signal to each of multiple mixer loads, which perform gaining and filtering of the corresponding mixed signal. The mixer further includes a second stage having gain stages each coupled to at least one of the mixer loads to weight the output of the corresponding mixer load and to provide an output to a summer. In an embodiment, each master RF device has a programmable weight (e.g., sine weighted) that can be based on a number of master RF devices to be activated.

Another aspect is directed to a multi-stage harmonic rejection mixer to receive and downconvert an incoming RF signal to a second frequency signal, which includes first and second stages. The first stage includes first and second sets of paths each to receive and process the incoming RF signal of a respective first and second polarity. In turn, each of the paths includes weighted transconductors having a weighting different than at least one other of the weighted transconductors to receive the incoming RF signal polarity and provide an RF current, a master LO device coupled to an output of the transconductor to receive and mix the RF current with a master clock signal to obtain a mixed signal, a rotating switch device coupled to the master LO device to cyclically switch the corresponding mixed signal to each of multiple mixer loads coupled to the rotating switch devices of the paths to perform gaining and filtering of the corresponding mixed signal. In turn, the second stage has a third set of paths each including a gain stage coupled to at least one of the mixer loads to weight the output of the corresponding mixer load and to provide an output to a summer.

A still further aspect is directed to a method for processing signals using a multi-stage rotating harmonic rejection mixer. The method includes passing an incoming RF signal through weighted transconductors to generate weighted RF signals, mixing the weighted RF signals with a master clock signal in mixer devices to generate weighted second frequency signals, cyclically rotating each of the weighted second frequency signals via a multi-phase bus to a mixer loads, outputting the weighted second frequency signals from each of the mixer loads to a corresponding gain stage, generating a double weighted second frequency signal in each of the gain stages, and combining the double weighted second frequency signals to output a second frequency signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a high level block diagram of a mixer in accordance with an embodiment of the present invention.

FIG. 2 is a block diagram of an implementation of a mixer in accordance with one embodiment of the present invention.

FIG. 3 is a schematic diagram of a rotating switch in accordance with an embodiment of the present invention.

FIG. 4 is a timing diagram of control signals for a rotating switch in accordance with one embodiment of the present invention.

FIG. 5 is a timing diagram of complementary control signals for a rotating switch in accordance with one embodiment of the present invention.

FIG. 6 is an example implementation of a gain stage of an intermediate frequency section in accordance with an embodiment of the present invention.

FIG. 7 is a schematic diagram of a quadrature mixer in accordance with an embodiment of the present invention.

FIG. 8 is a block diagram of a portion of a rotating harmonic rejection mixer in accordance with an embodiment of the present invention.

FIG. 9 is a timing diagram of an IF current of a single IF stage in accordance with one embodiment of the present invention.

FIG. 10 is the timing diagram of FIG. 9 in the presence of mismatches in accordance with an embodiment of the present invention.

FIG. 11 is a timing diagram of an IF current of a single IF stage in the presence of noise in accordance with an embodiment of the present invention.

FIG. 12 is a block diagram of a system in accordance with one embodiment of the present invention.

FIG. 13 is a flow diagram of a method in accordance with an embodiment of the present invention.

FIG. 14 is a block diagram of a two-stage mixer in accordance with one embodiment of the present invention.

FIG. 15 is a block diagram of a single path of the RF portion of the multi-stage mixer in accordance with one embodiment of the present invention.

FIG. 16 is a timing diagram of a series of pulses output by a shift register in accordance with one embodiment of the present invention.

FIG. 17 is a block diagram of a pulse generation circuit in accordance with an embodiment of the present invention.

FIG. 18 is a block diagram of an arrangement of an RF section of a multi-stage mixer in accordance with an embodiment of the present invention.

FIG. 19 is a block diagram of an IF section of a mixer in accordance with an embodiment of the present invention.

DETAILED DESCRIPTION

In various embodiments, a rotating harmonic rejection mixer may be provided to enable improved harmonic rejection for a mixing operation between an incoming radio frequency (RF) signal and a clock frequency signal, such as a master clock which may be a square wave signal whose frequency is a multiplied version of a local oscillator (LO) frequency. In various embodiments, the rotating harmonic rejection mixer may be controlled to enable some or all of a plurality of gain stages of an intermediate frequency (IF), which are then summed to provide an output IF signal for further processing in a given receiver. As used herein, the term “rotating” means that an output of a mixing operation is cyclically rotated to different IF gain stages during a given time period.

Due to the design of the mixer in which the incoming RF signal is downmixed, e.g., to an IF frequency, after which this IF signal is processed by way of gaining, filtering and so forth, any mismatch-causing devices do not operate at high frequencies. Accordingly, better matching passive components may be used and feedback around active devices is also implemented to improve harmonic rejection at significantly lower power and area consumption. Furthermore, in various implementations a mixer may include a single RF device, e.g., a single differential transconductor controlled by a single switching pair. In these implementations, because there is only a single RF device, no mismatches occur in the RF section.

Furthermore, these front-end devices, i.e., in the RF portion and a local oscillator path, can be formed of minimal-sized devices leading to improved power and area reductions. Also, by using such smaller size devices, bandwidth of a front-end amplifier that provides signals to the mixer, such as a low noise amplifier (LNA) may be increased, as mixer input capacitance can be significantly reduced. In this way, LNA power dissipation may be reduced while providing more flexibility for selection and design of an LNA. In some implementations, resistor areas in an IF section may be optimized such that resistors implementing the peak of a sine wave implementing the phases of the individual IF portions can be wider than those at the rising portion of the sine wave.

Instead of a single front-end device in other embodiments a multi-stage harmonic rejection mixer may be provided. In these implementations, multiple paths each to receive a single incoming RF signal may be present. Furthermore, each of these paths may include multiple weighted elements, e.g., according to a sine function to thus spread gain across multiple elements of these paths. In this way, any gain errors that arise can be reduced, owing to a product of the gain errors inherent in these various weighted devices. For example, in some embodiments both front-end devices, namely multiple transconductors, may be weighted according to a first sine function, and IF devices, namely gain devices such as resistors, can be weighted according to a second sine function. In some implementations, such as where an equal number of RF paths and IF paths are present, these first and second sine functions may be proportional to each other, although the scope of the present invention is not limited in this regard.

Note that in an implementation incorporating a multi-stage harmonic rejection mixer, the rotating switch devices of each of the paths may be switchably connected to each of the multiple IF paths, e.g., via a multi-phase IF bus, to thus provide rotating and cyclic switch connections in accordance with an embodiment of the present invention.

While the scope of the present invention is not limited in this regard, such a mixer may be incorporated into various receivers such as a television receiver, radio receiver or other receiver of incoming RF signals. Because the number of such gain stages can be dynamically controlled, embodiments may provide for control of an amount of harmonic rejection to be provided, which may vary given a frequency at which the incoming signals are received. For example, in the context of a television receiver, incoming signals may be received via broadcast of over-the-air signals at VHF or UHF frequencies or via broadband cable at a higher frequency. Depending upon the frequency at which the tuner operates, differing amounts of gain stages may be provided to enable a controllable amount of harmonic rejection to be realized, while also preventing flicker noise in the mixer. Furthermore, by reducing the number of gain stages enabled during operation at certain frequencies, reduced power consumption may be realized.

Referring now to FIG. 1, shown is a block diagram of a mixer system in accordance with an embodiment of the present invention. As shown in FIG. 1, mixer system 10 is coupled to receive an incoming RF signal at a mixer 20, which mixes the incoming signal with a master clock signal. The master clock may be at a frequency of N×LO, where N is an integer corresponding to a number of gain stages (discussed below) in mixer system 10 and LO is at an output frequency of an LO (not shown in FIG. 1). In various embodiments, the LO may include a voltage controlled oscillator (VCO) that generates a sine wave, that in turn may be modified into a square wave signal (e.g., via a frequency divider) along an LO path such that the master clock provided to mixer 20 is a square wave signal. Mixer 20 may thus multiply the incoming RF signal by the master clock and the result is then rotated between a plurality of individual gain stages 30 ₀-30 _(N−1) (generically gain stage 30). As will be described further below, each gain stage 30 has a different gain factor a₀-a_(N−1) associated therewith.

A switch 25 may be controlled to cyclically rotate the output from mixer 20 to each of gain stages 30. The angular velocity of rotation sets the effective LO frequency. For example, if switch 25 has completed one rotation in N cycles of the master clock, the effective LO equals the master clock frequency divided by N. In various implementations, switch 25 may be controlled to be connected to a given gain stage 30 when the LO is at a high value (i.e., when there is a signal through switch 25). When there is no signal through switch 25 (i.e., when the LO is at a low state), it may be rotated to the next gain stage 30. In this way, switch 25 does not contribute any noise, and any offsets within operation of switch 25 do not contribute to any harmonic rejection degradation.

Referring still to FIG. 1, each gain stage 30 has an input IF port (IF₀-IF_(N−1)) that is coupled to receive the output from mixer 20 when switch 25 enables interconnection to the given IF port. Each gain stage 30 scales signal at the IF ports (IF0 to IFn−1) by a different factor, or degree, to produce a resultant signal that is provided to a summer block 40 that combines the outputs of all such gain stages 30 to generate an IF output which may be provided to further circuitry of a tuner to process the received signal. Note that in FIG. 1, IF filters, which may be also referred to herein as mixer loads, that are present on each of the IF ports to enable gaining, filtering and conversion of current to voltage, and coupled between switch 25 and a corresponding gain stage, are not shown for ease of illustration.

Each gain stage 30 may have a different coefficient a₀-a_(N−1) that may be selected to cancel harmonics in the incoming signal. More specifically, in some embodiments the a_(k) coefficients where k equals zero to N−1 may be selected based on the following periodic function of the square wave phase:

$\begin{matrix} {a_{k} = {{\sin\left( {\frac{2\pi}{N}k} \right)}.}} & \lbrack 1\rbrack \end{matrix}$

By selecting a given value of N, the harmonics that are cancelled by mixer system 10 may be controlled. As described above, the summation of all of the individual gain blocks 30 (i.e., phases) may be summed at summer block 40 and provided to additional receiver circuitry.

Implementations of a mixer to enable controllable harmonic rejection as well as noise immunity may take different forms. Referring now to FIG. 2, shown is a block diagram of an implementation of a mixer in accordance with one embodiment of the present invention. As shown in FIG. 2, mixer 100 is one example implementation in which an incoming RF signal having positive and negative components RF_(N) and RF_(P) are provided to master RF devices 102. As shown in FIG. 2, master RF devices 102 may include n-channel metal oxide semiconductor field effect transistors (nMOSFETs) 105 and 106 which act as transconductors having gate terminals connected to RF signals RF_(P) and RF_(N), respectively, and providing an RF current through a drain terminal. Thus only a single differential transconductor is present, avoiding any mismatch.

The outputs of master RF devices 102 are provided to master LO devices 110. Master LO devices 110 may act to mix the incoming RF signal with the master clock frequency. Specifically, as shown in FIG. 2 master LO devices 110 may include a plurality of nMOSFETs 112-118 having source terminals coupled to receive an RF output from master RF devices 102 and gate terminals coupled to receive the master clock signal (respectively N×LO_(P) and N×LO_(N)) and having drain terminals to provide an IF output to a plurality of rotating switches 120-126 (generically rotating switch 120). Rotating switches 120 may be controlled by control signals, LO_(a+b)<N−1:0> (note LO_(a) and LO_(b) are distinct signals) respectively, to cyclically switch the output from the master clock devices to each of a plurality of IF output ports, IF<N−1:0>, that in turn are provided to a plurality of IF input ports (e.g., IF_(N−1)-IF₀) of a plurality of mixer loads 130-136 (generically mixer load 130).

Note that the rotating switches in the mixer processing the negative RF signal, namely RF_(n), are connected differently to the IF ports, IF₀ to IF_(n−1). Since RF_(n) is 180 degrees phase shifted with respect its counterpart, RF_(p), the rotating switches' outputs should also be phase shifted by 180 degrees. Since a rotation through N stages implies a phase shift of 360 degrees, a phase shift of 180 degrees is obtained by cyclically shifting through N/2 stages. Thus if a rotating switch in the mixer processing RF_(p) was connected to the IF port, IF_(k), then the corresponding rotating switch in the mixer processing RF_(n) would be connected to IF port IF(_(k+N/2)) or IF_((k−N/2)) depending on whether k was lesser than N/2 or not, respectively (thus the arrayed output of the rotating switches connected to mixer processing RF_(n) is represented in FIG. 2 as IF<N/2−1:0>, (followed by), IF<N−1,N/2> whereas the arrayed output of the rotating switches connected to mixer processing RFp is represented as IF<N−1,0>.) Mixer loads 130 may perform gaining and filtering of the IF signals, and may be RC filters in some embodiments. In some implementations, each such mixer load 130 may have the same RC weighting. The output of the respective mixer loads 130 may be provided through additional gain stages that may include buffers and impedances, and then on to summing blocks (not shown in FIG. 2), which sum the respective phases and provide the IF output to a desired location.

Rotating switches such as switches 120 shown in FIG. 2 can be implemented in a variety of manners. Referring now to FIG. 3, shown is a schematic diagram of an example implementation for a rotating switch in accordance with an embodiment of the present invention. As shown in FIG. 3, rotating switch 120 may include a plurality of nMOSFETS 121 _(N)-121 ₀ (generically MOSFET 121). As shown, each MOSFET 121 has a source terminal coupled to receive a current input from a master LO device 110. Further, each MOSFET 121 may have a gate terminal controlled by a different one of a plurality of control signals LO<0:N−1>. As described above, each MOSFET 121 may be cyclically controlled to enable each MOSFET 121 to output via its drain terminal the current input signal for a given portion of the LO cycle. Thus the outputs of rotating switch 120 may be provided at a plurality of output IF ports IF<0:N−1> that in turn couple to corresponding IF input ports of, e.g., mixer loads 130 of FIG. 2.

To enable rotating switch 120 to rotate the input current between its various outputs, the gates of MOSFETs 121 may be driven in accordance with the timing diagram shown in FIG. 4, in one embodiment. FIG. 4 shows the master clock frequency for the positive input current (i.e., N×LO_(P)) which is a square wave signal. As shown in FIG. 4, for a given period of the LO, N cycles of the master clock may be generated. In turn, each MOSFET of the rotating switches may be controlled cyclically by a given control signal LO_(a)<N−1:0>. Note that these control signals are also square wave signals that may have a pulse width equal to about a time period of the master clock. Furthermore, note that the control signals are individually enabled to a high state during a low portion of the master clock. While shown as switching at approximately halfway through a low state of the master clock frequency, the scope of the present invention is not limited in this regard. By enabling each MOSFET in turn, during a single period of the LO each MOSFET may be enabled for a time period of LO divided by N.

As shown in FIG. 5, a similar timing mechanism may be enabled for the negative master clock signals, i.e., N×LO_(N). Note that the positive and negative master clock frequencies are complementary versions of each other. The transitions in the control signals LO_(a)<K> and LO_(b)<K>, where K equals 0 to N−1 occur when the master clock is at a low state. Thus there is no current in the rotating switches when the gates of the MOSFETs are being toggled, and any random offsets between the MOSFETs in the rotating switches do not affect their output. Note that the same is true for any noise in the switches. Still further, when the master clock is at a high level, the MOSFETs of the rotating switches simply act as cascodes and thus their noise is not important. As a result, any mismatches in the master LO devices affect all outputs of the rotating switches equally. Still further, any mismatches in the master RF devices affect all outputs of the rotating switches equally, because it is the same RF current that is cyclically rotated between the different mixer outputs. Accordingly, mismatches in both the RF and LO devices do not cause any harmonic rejection degradation. Instead, the only components that could cause such harmonic rejection degradation are devices in the IF portions (i.e., mixer loads 130 of FIG. 2). For a downconversion mixer such as used in various receiver architectures, because the IF is at a much lower frequency than the received RF signal, negative feedback stages may be provided and any gain/phase errors caused by these stages are largely dependent on passive components such as resistors and capacitors, and not active components such as transistors. Because for a typical IC process such as a CMOS process, passive components match much better than their active counterparts, enabling mismatch errors between passive components can greatly reduce harmonic rejection degradation.

A mixer in accordance with an embodiment of the present invention thus shifts the device matching problem of harmonic rejection from high frequency RF/LO devices to lower frequency IF devices, and further shifts device matching issues from poorly matching active devices to better matching passive devices. Such a mixer can achieve improved harmonic rejection while reducing both power consumption and die area consumed by the mixer.

As described above, in some embodiments mixer loads 130 of FIG. 2 may be implemented as RC filters. Referring now to FIG. 6, shown is an example implementation of a gain stage of an IF section in accordance with an embodiment of the present invention. As shown in FIG. 6, IF section 200 includes a plurality of IF input ports, IF<N−1>-IF<0>, to receive the output of the master RF and LO devices and mixer loads of FIG. 2 (generically 202). Each input port is coupled to a respective unity gain buffer 210 _(N−1)-210 ₀. While the scope of the present invention is not limited in this regard, in one embodiment a unity gain buffer may be implemented with a pair of MOSFETs, namely a pMOSFET coupled to receive incoming voltage at a gate terminal and having source and drain terminals coupled to different current sources, respectively. In turn, a second MOSFET, which may be a nMOSFET, may have a gate terminal coupled to a drain terminal of the first MOSFET, a source terminal coupled to ground, and a drain terminal coupled to an output terminal (also coupled to the source terminal of the first MOSFET). Such a gain buffer may have a gain set in accordance with the following equation:

$\begin{matrix} {\frac{Vout}{Vin} = \frac{A}{1 + A}} & \lbrack 2\rbrack \end{matrix}$ where A is the open loop gain of unity gain buffer. Note that a large value for A may help reduce mismatch in gains.

Referring still to FIG. 6, unity gain buffers 210 are each coupled to a corresponding resistance 220 _(N−1)-220 ₀. In various embodiments, each resistance 220 may be of a different value. More specifically, in one embodiment:

$\begin{matrix} {{{R_{I}\left\langle k \right\rangle} = \frac{Runit}{\sin\left( {\frac{2\pi}{N}k} \right)}}{{{for}\mspace{14mu} k} = {{0\mspace{14mu}{to}\mspace{14mu} N} - 1}}} & \lbrack 3\rbrack \end{matrix}$ where Runit is a unitary or normalized resistance value. In some implementations, the unitary resistance value may be based on a resistor ratio such that the different resistors approximate a sine wave as closely as possible to improve harmonic rejection. In some embodiments, the integer ratios may be implemented with resistors connected in parallel for each of R₁<k>, with each resistor of a uniform length/width. These resistor ratios may be integer approximations of a sine wave in some embodiments. For example, in one implementation for 16 sine wave coefficients, a plurality of integer values may be chosen to provide for third order harmonic rejection of approximately 56 dB, with fifth order harmonic rejection of approximately 53 dB and seventh order harmonic rejection of approximately 48 db. As shown in Table 1 below, various integer fits for a sine wave may be used in different embodiments, which provide for various levels of harmonic rejection, in one implementation.

TABLE 1 dB dB dB dB Sine wave coefficient (N = 16) S/3 S/5 S/7 S/9 9 17 22 24 22 17 9 0 56 53 48 51 2 12 20 25 26 23 17 8 47 56 42 54 8 17 23 26 25 20 12 2 47 56 52 54 3 13 21 26 27 24 17 8 52 50 48 51 8 17 24 27 26 21 13 3 52 50 48 51 4 14 22 27 28 24 17 7 45 46 51 53 4 15 24 29 30 26 18 8 51 49 48 51 8 18 26 30 29 24 15 4 51 49 49 51

Furthermore, if quarter sine wave coefficients are used, integer ratios of 0, 9^(1/5), 17, 22^(1/5), and 24 may be realized for a quarter sine wave, providing harmonic rejection in excess of 65 dB. By using integers to approximate a sine wave, immunity from end effects and modeling errors may be realized. Note further that the different weighting values used may be applied in different order to the phases than that shown above.

Still referring to FIG. 6, the outputs of each resistance 220 is provided to a differential amplifier 240, which acts to sum all of the signals and provide a differential voltage output I_(output) at a desired IF frequency. Note that feedback resistors, R_(feedback), are coupled between the respective output and input terminals of differential amplifier 240. While shown with this particular implementation in the embodiment of FIG. 6, the scope of the present invention is not limited in this regard. For example, in other implementations rather than the unity gain buffers followed by resistors of FIG. 6, transconductors may be used instead.

FIG. 6 thus shows an implementation that provides a differential IF signal. In other implementations, quadrature signals may be obtained using a different implementation by providing another set of unity gain buffers that in turn are coupled to another set of resistances whose values mimic a “cosine” wave rather than a “sine” wave. In such an implementation, the quadrature phase resistances may be in accordance with the following equation:

$\begin{matrix} {{{R_{Q}\left\langle k \right\rangle} = \frac{Runit}{\cos\left( {\frac{2\pi}{N}k} \right)}}{{{for}\mspace{14mu} k} = {{0\mspace{14mu}{to}\mspace{14mu} N} - 1}}} & \lbrack 4\rbrack \end{matrix}$

Referring now to FIG. 7, shown is a schematic diagram of a quadrature mixer in accordance with an embodiment of the present invention. As shown in FIG. 7, mixer 300 may be coupled to receive an incoming RF signal (i.e., RF_(N) and RF_(P)) and positive and negative master clocks (i.e., N×LO_(N) and N×LO_(P)) and mix these signals in a mixer portion 310 which then provides IF outputs to an in-phase IF portion 320 and a quadrature-phase IF portion 330. As described above, in-phase IF portion 320 may include a plurality of unity gain buffers 322 _(N−1)-322 ₀ that in turn are coupled to resistances 324 _(N−1)-324 ₀, outputs of which in turn are provided to a differential amplifier 325. Similarly, quadrature in-phase IF portion 330 may include a plurality of unity gain buffers 332 _(N−1)-332 ₀ that in turn are coupled to resistances 334 _(N−1)-334 ₀, outputs of which are provided to a differential amplifier 335. In this way, mixer 300 provides I and Q IF outputs.

In various embodiments, improved image rejection may be realized in a quadrature mixer. This is so, as matching between I and Q outputs is solely determined by matching in the IF section, for the same reasons discussed above. That is, because mismatches in the master LO devices and master RF devices do not cause any gain/phase errors between the different IF<k> outputs, the quadrature signals derived from these IF outputs have improved image rejection.

In addition to improved harmonic and image rejection provided by embodiments of the present invention, better second-order intermodulation products (IP₂) also can be achieved.

Referring now to FIG. 8, shown is a block diagram showing a portion of a rotating harmonic rejection mixer in accordance with an embodiment of the present invention. Note mixer 100 of FIG. 8 may correspond to the positive portion of mixer 100 shown in FIG. 2. Thus, an incoming RF signal is passed through a transconductor 105, providing positive and negative current portions I_(P) and I_(N) through MOSFETs 112 and 114. In turn, these currents are switched through rotating switches 120 and 122. Only a single one of the switch outputs is shown, namely IF₀, from rotating switches 120 and 122. This current I_(IF)<0> is thus a combination of the individual positive and negative currents from rotating switches 120 and 122.

Referring now to FIG. 9, shown is a time domain analysis of the IF current provided by such a single one of the IF output ports (i.e., I_(IF)<k> for any k), which is a combination of both positive and negative currents provided by master LO devices 112 and 114 for the duration of one N×LO time period. As shown in FIG. 9, both the positive and negative input currents (Ip and In) are square waves that complement each other such that the combined current into any IF port has a pulse width equal to the sum of pulse widths of Ip and In. Thus, the pulse width of the current pulse into any IF port is equal to one period of N×LO clock.

In the presence of mismatches in the active devices, e.g., in the master LO devices, the time domain of FIG. 9 may evolve into the time domain of FIG. 10, in which the dashed lines show the corresponding waveforms in the presence of mismatches in the active components of the master LO devices. Because of the complementary nature of the signals, mismatches in these master LO devices do not change the pulse width of the IF output current, I_(IF)<k> for any k, as shown in FIG. 10. Accordingly, the low frequency IM₂ component appears the same way in each of the IF outputs and is ultimately cancelled in the IF stage as:

$\begin{matrix} {{IF} = {\sum\limits_{K = 0}^{N - 1}\;{a_{K}x\mspace{14mu}{IF}\mspace{14mu}\left\langle k \right\rangle}}} & \lbrack 5\rbrack \end{matrix}$ where

$a_{K} = {\sin\left( \frac{2\pi\; k}{N} \right)}$ for k=U to N−1, because for every k, a_(K)=−a_(K+N/2). Accordingly, mismatches in the master LO devices do not contribute to IP₂ degradation and instead IP₂ degradation is solely determined by low frequency matching in the IF section, similar to that described above with regard to harmonic rejection degradation.

In addition to mismatches that may exist between the MOSFETs of the master LO devices, flicker noise may also be present. Referring now to FIG. 11, shown is a timing diagram of a current into a given IF port IF<k>, which is obtained by summing the positive and negative currents I_(N) and I_(P). As shown in FIG. 11, the dashed lines indicate current waveforms in the presence of flicker noise in the MOSFETs of the master LO devices. Note that the pulse width of IF<k> and thus the DC component is unaffected by this flicker noise. Accordingly, even in the single-ended outputs of IF<k>, flicker noise of the LO switching devices is absent. Accordingly, there is no need to rely on matching between positive and negative sides to obtain differential low flicker noise.

Still further, embodiments of the present invention provide for lower input referred thermal noise as compared to a conventional square wave mixer. That is, given the same total transconductor current, input referred noise of a mixer in accordance with an embodiment of the present invention may be significantly smaller, for example, on the order of 2(π²/8), as noise downconversions from LO harmonics which may be present in a conventional mixer are absent in embodiments of the present invention.

Embodiments may be implemented in many different system types. As described above, applications may include mixed signal circuits that include both analog and digital circuitry. Referring now to FIG. 12, shown is a block diagram of a system in accordance with one embodiment of the present invention. As shown in FIG. 12, system 1000 may be a television that is coupled to receive a RF signal from an antenna source, cable distribution, or other source. The incoming RF signal may be provided to a television tuner 1005 which may be, in one embodiment a single-chip mixed signal device. Television tuner 1005 may incorporate embodiments of the present invention to provide improved harmonic and image rejection while consuming lower power and area.

Referring still to FIG. 12, tuner 1005 includes a bandpass filter 1110 having an output coupled to a low noise amplifier (LNA) 1115 to receive and amplify the RF signal from an antenna 1001. The output of LNA 1115 is provided to another bandpass filter 1120 that in turn is coupled to mixer 1125, which may be a rotating mixer in accordance with an embodiment of the present invention. As shown in FIG. 12, mixer 1125 receives a master clock signal having a frequency of N×LO from a frequency divider 1135 which in turn is coupled to receive a VCO frequency from a voltage control oscillator (VCO) 1130. While the scope of the present invention is not limited in this regard, the VCO frequency may be at a relatively high frequency, e.g., at least several GHz. In turn, frequency divider 1135 generates a divided frequency that is provided at the master clock value, which may be at a frequency of between approximately 1000 and 2000 MHz. In turn, mixer 1125 downconverts the incoming RF signal with the master clock signal to generate a complex IF output at a frequency of less than approximately 10 MHz. The complex I and Q IF signals output from mixer 1125 are provided to unity gain buffers 1140 a and 1140 b and are filtered by lowpass filters 1145 a and 1145 b and then may be digitized by ADCs 1150 a and 1150 b.

Referring still to FIG. 12, the digitized output of tuner 1005 may be provided to additional processing circuitry within television 1000, such as a demodulator 1170 and associated circuitry to enable a processed television signal to be provided to a display 1175. While shown with this particular implementation in the embodiment of FIG. 12, it is to be understood the scope of the present invention is not limited in this regard. Furthermore, it is to be understood that embodiments may be implemented in many different devices, such as receivers, transmitters and so forth. Still further, control logic, program storage or other computer readable media may be present to store instructions that when executed within a processor of tuner 1000 perform control of a number of gain stages, master clock frequencies, and thus harmonic rejection provided.

Referring now to FIG. 13, shown is a flow diagram of a method in accordance with an embodiment of the present invention. As shown in FIG. 13, method 500 may be used to dynamically adjust a number of gain stages in a mixer. By doing this, reduced power consumption may be realized while achieving a desired level of harmonic rejection, image rejection and so forth for incoming signals of a given frequency. As shown in FIG. 13, method 500 may begin by receiving a request for a channel at a given frequency (block 510). For example, a tuner that is part of a television receiver may receive a request from a user for programming on a certain channel. Based on that frequency, which is at a given frequency based on the band in which the channel is included, the number of stages N to be enabled in a mixer may be determined (block 520). For example, the following Table 2 shows the number of stages in the mixer (N) for different input frequency ranges. Having different values of N for different input frequency ranges enables a mixer to reject only desired number of harmonics while saving power in the front end dividers (e.g., 1135 in FIG. 12). The value of N may be controlled by programming of software, e.g., during incorporation of a tuner into a given system.

TABLE 2 Input Frequency Range Mixer stages (N)  40 MHz-125 MHz 16 105 MHz-170 MHz 12 155 MHz-255 MHz 8 210 MHz-337 MHz 6

Referring still to FIG. 13, then a master clock frequency may be selected based on this determined number of stages N (block 530). That is, a LO frequency may be provided to a clock multiplier, frequency divider or so forth that generates a master clock frequency of N.LO. This master clock frequency may thus be provided to the rotating mixer (block 540). Using this signal and an incoming RF signal that includes the desired channel, a mixing operation may occur such that a mixed signal corresponding to an IF signal may be switched to each of the N IF stages during each master clock cycle (block 550). The sum of these IF outputs for each clock cycle may be summed and sent along for further processing (block 560). While shown with this particular implementation in the embodiment of FIG. 13, the scope of the present invention is not limited in this regard. As described above, in some embodiments machine-readable instructions may implement the method of FIG. 13.

As discussed above, in some embodiments a harmonic rejection mixer can be implemented using a two-stage architecture. In this way, gain errors associated with the mixer can be reduced, as a resulting product of gain errors of the first and second stages of the mixer reduces the total gain error. Referring now to FIG. 14, shown is a block diagram of a two-stage mixer in accordance with one embodiment of the present invention. As shown in FIG. 14, mixer 400 is a two-stage mixer including a first stage 401 and a second stage 402. As seen, first stage 401 includes multiple paths, including a first set of paths 403 _(a) to receive an incoming RF signal of a first polarity and a second set of paths 403 _(b) to receive the incoming RF signal of a second polarity. Second stage 402 may also include a set of paths 403 _(c) to receive and further process IF signals received from the first and second sets of paths. In different embodiments, differing numbers of paths may be present in each of these sets (or enabled, depending on frequency of the desired channel). For example, in one embodiment a total of N paths may be provided, with N/2 paths present in each of the first and second sets of paths and N paths in the third set of paths. However, understand the scope of the present invention is not limited in this regard in varying amounts of paths may be present in the first and second stages. Furthermore, while described herein with regard to a two-stage architecture, understand the scope of the present invention is not limited in this regard and in other implementations a X-stage mixer may be present, where X is greater than 2.

Details of a representative path of mixer 400 of will be described. Specifically, a first path includes a RF device 410 ₀ which may correspond to a master RF device that is coupled to receive the incoming RF signal RF_(p) corresponding to a positive portion of the RF signal. In turn, master RF device 410 ₀, which may be implemented with a single transconductor, may be coupled to a mixer 420 ₀. Similar to that discussed above, mixer 420 ₀ may mix the RF signal with a clock signal corresponding to a master clock signal, e.g., having a value of N×LO. Accordingly, mixer 420 ₀ thus downconverts the incoming RF signal to a lower frequency, e.g., an IF frequency. This IF frequency signal may be provided to a rotating switch device 425 ₀. Rotating switch device 425 ₀ may cyclically switch the product of the RF signal with the master clock to a plurality of load devices 428 ₀-428 _(N−1). For each cycle of the master clock signal, rotating switch 425 ₀ may switch the IF signal to one of the load devices, which as seen each may be formed of a respective RC circuit. The processed signals from each of these paths may then be passed to the second stage 402, and more specifically through a corresponding gain device, e.g., 430 ₀, which in one embodiment may include weighted resistors as described further below. From there, the gained and filtered signal may be provided to a summer 440.

During operation at any given time instant, each one of switches 425 ₀-425 _(N−1) may be coupled to a distinct one of load devices 428 ₀-428 _(N−1). For example, with reference to Table 3, if at some time period the switches are connected as in the middle column of Table 3, then at a next period, the switching may be connected as in the right column of the table.

TABLE 3 Time = t Time = t + T_(LO/N) S₀ → IF_(K) IF_(K+1) S₁ → IF_(K+1) IF_(K+2) . . . . . . . . . S_(N−1) → IF_(N−K) IF_(N−K+1)

Thus for each cycle of the master clock signal, each one of these switches may enable coupling of the IF signal of the corresponding path to be provided to one of the load devices. Then during each low or null value of the master clock signal, all of the rotating switches of the N paths may switch to cause its corresponding IF signal to be passed to a successive one of the load devices. Thus all switches S₀ to S_(N−1) are rotated together from IF_(K)-IF_(N−1) to IF_(K+1)-IF_(N−K+1) and so on. Rotation accordingly occurs when there is no current through the switches.

As further seen in FIG. 14, note that each of the master RF devices may be weighted, e.g., according to a sine weighting 405. The actual weights for each of the master RF devices can be based on the number of present/active paths. For example, in one embodiment, the weighting of each transconductor Gm_(k) (where k=0 to N−1) may be proportional to sin (2πk/N). To provide for such programmable weighting of the gain paths, the transconductors themselves may have a programmable size to accommodate the weighting. Also, a similar weighting may be provided to the gain devices 430 of the mixer. That is, gain devices 430 may also have a sine weighting also dependent on the number of present/active stages. In one embodiment, the resistor weightings R_(k) for the gain devices of each path may be proportional to 1/sin(2πk/N). Note that the actual weightings of the transconductors and the gain stages need not be the same, and instead can be proportional to each other. In yet other embodiments, there may be no proportionality between the different weightings, particularly in embodiments in which there are differing numbers of transconductors and gain stages.

Referring now to FIG. 15, shown is a block diagram of a single path of a portion of the RF portion of the multi-stage mixer. As seen in FIG. 15, a given path may include the master Gm device 610, and that in turn is coupled to a mixer device 620 that mixes the RF signal with a master clock signal of N times the LO frequency. In turn, this IF signal may be provided to a rotational switch 625. Accordingly, a single polarity RF signal may be downconverted and output from rotational switch 625 as a plurality of IF signals according to a cyclical rotation in accordance with an embodiment of the present invention.

More specifically with regard to FIG. 15, note that an incoming single-ended RF signal is provided to a gate of a master Gm device 610. As seen, master Gm device 610 may be formed of a transconductor M1 having a gate terminal coupled to receive this incoming RF signal, a source terminal coupled to an impedance, and a drain terminal coupled to mixer stage 620. With regard to mixer stage 620, note that the incoming RF signal is provided to source terminals of a pair of MOSFETs M2 and M3, each of which may be gated by a corresponding polarity of the master clock signal MLO, in turn providing at a drain terminal of the corresponding MOSFET the mixed product. The mixed products are in turn provided to rotational switch 625 that in turn is formed of a plurality of individual switches 621 ₀-621 _(N−1) and 623 ₀-623 _(N−1). These individuals switches may be configured as MOSFETs having source terminals coupled to receive the mixed product of the RF signal with master clock, and gate terminals coupled to receive a corresponding control signal. More specifically, as seen in FIG. 15, rotational switch 625 may include (or may be coupled to) a shift register 624 that has N stages. An incoming pulse signal is provided to an input of shift register 624 and is clocked by the corresponding polarity of the master clock signal to thus shift the pulse value through the shift register. As seen, each stage may have an output port that in turn is coupled to a gate terminal of one of the individual switches 621/623. The pulse width of each of these outputs from shift register 624 may be at a period of the master clock signal, or T_(LO)/N.

Note that the shift register is clocked by the master clock signal MLO and in turn generates a series of pulses each having a pulse width corresponding to the period of the master clock signal. Note further that the switching of the pulses occurs during a low portion of the master clock signal. Accordingly, at the rising edge of each of these pulses, a given switch is activated to provide a path through that switch from mixer stage 620 on a given IF output signal line from rotational switch 625. As time advances, the rotation continues through the N IF outputs from rotational switch 625. Note that while illustrated in FIG. 15 as including two shift registers 624, understand that in various embodiments, a single shift register may be provided, with each stage providing two outputs, one output for a corresponding one of switches 621 and the other output for a corresponding one of switches 623. While the scope of the present invention is not limited in this regard, in some embodiments the pulse signal provided as the input to shift register 624 may in turn be generated by a pulse generation circuit that itself can be clocked by the master clock signal.

Referring now to FIG. 17, shown is a block diagram of a pulse generation circuit in accordance with an embodiment of the present invention. Various approaches to the generation of rotational pulses may be used, including a Grey coded counter and decoder, a shift register with set/reset, or a counter followed by a shift register. In one embodiment, the counter approach may be chosen primarily because of its low power nature and relative simplicity. As shown in FIG. 17, pulse generator 800 may generate the rotational pulses used to provide the different phases of mixer outputs to the IF network. As seen in the embodiment of FIG. 17, a shift register 810, which may be an N-stage shift register receives a clock signal (CLK), which may be at the MLO, and a pulse output from a pulse generator circuit 820. Each stage of the shift register may generate rotational pulse signals on a falling edge of the clock. In one embodiment, the N-stage shift register can be made of N master-slave D flip-flops. The signals P_(O)-P_(n−1) (a timing diagram for which is shown in FIG. 16) are generated from the falling edge of the clock, and the pulse widths of each pulse Px is equal to one time period of the incoming clock, thus satisfying the requirements for rotational pulses of the RHRM. As seen, pulse generator circuit 820 may be formed of a series set of a Count-till-N (block 825) and D type flip-flops, including a first flip-flop 830 coupled to a second flip-flop 840. The first flop 830 retimes the output of the count-till-N block and the second flop 840 delays the inverted output of 830. As seen in FIG. 17, the Q outputs of the first and second flip-flops may be logically combined in a NAND gate 845 to generate a pulse signal, as shown in FIG. 17. While shown with this particular implementation in the embodiment of FIG. 17, understand that the scope of the present invention is not limited in this regard.

Referring now to FIG. 18, shown is a block diagram of an arrangement of an RF section of a multi-stage mixer in accordance with an embodiment of the present invention. As shown in FIG. 18, mixer 950 includes a plurality of master RF/mixer stages 960 ₀-960 _(N). In general, each master RF/mixer stage 960 may include a master transconductor and mixer device such as shown in FIG. 15 at reference numerals 610 and 620. Accordingly, the output of each of these master RF/mixer stages is the mixed product of the RF signal with the master clock that in turn is provided to a corresponding rotational switch stage 970 ₀-970 _(N−1). As seen, each stage 970 receives one of the pulse outputs P_(o)-P_(n−1) from shift register 810 of FIG. 17 and according to the timing diagram of FIG. 16. Each of these switch stages cyclically and rotatingly switches the output from stage 960 to a plurality of IF output lines, IF₀-IF_(N−1), which may be part of a multi-phase IF bus 975 that in turn is provided to a load network 980 which may be formed, in one embodiment, as a plurality of RC filters. Although not shown for ease of illustration in FIG. 18, understand that the outputs of these load filters may be provided via the multi-phase IF bus to unity gain buffers, outputs of which may in turn be provided to programmably weighted resisters that in turn can be coupled to one or more summing amplifiers. In one embodiment, the unity gain buffers and resistors, along with the summing amplifiers, may correspond to the second stage of the mixer, and thus with reference back to FIG. 14, may correspond to gain devices 430 and summer 440.

FIG. 19 is a block diagram of an IF section of a multi-stage mixer in accordance with an embodiment of the present invention. The IF stage may operate to further weight the multi-phase signals IF₀ to IF_(N−1) generated by the RF section of the mixer. In one embodiment the weighting occurring in the IF section may be proportional to the weightings of the RF section. In a particular implementation, each of the IF paths may be weighted according to sine wave coefficients, and then the outputs can be summed to generate the final output corresponding to a downmixed signal, which in various embodiments may be at an IF frequency and in quadrature form, i.e., as I_(out) and Q_(out). As seen in FIG. 19, the IF section 1100 includes unity gain buffers 1110 coupled to receive multiple IF phases via a multi-phase IF bus and to output buffered IF phases via the multi-phase IF bus. In turn the buffered IF phases may be provided to a plurality of weighting resistors 1130, 1140 and operational amplifiers 1150 _(a) and 1150 _(b) each having an RC network coupled across it. These operational amplifiers may act to sum the inputs through each set of weighting resistors to thus obtain a differential quadrature output. Thus the IF outputs from each of the RF paths (e.g., of FIG. 18) may be provided to through buffers 1110 to corresponding cosine weighted resistors 1130, while in turn the IF outputs may first pass through a phase shifter 1135 before being provided to sine weighted resistors 1140. The unity gain buffers can be reused for different numbers of paths, leading to area savings, and the resistors can be programmable for different paths. In one implementation the sine wave coefficient weightings can be accomplished through resistors 1130 and 1140, which are implemented as an integer number of parallel unit resistors. For example, for a N-weight unit, N parallel resistors of the unit value may be present. This approach makes the relative ratio of any two sine wave coefficients exact and immune to modeling deficiencies. The fractional part of the weights can be implemented by using the unit resistor in series. For example, a plurality of unit resistors (a number of which depends on desired weights) can be provided and which are controlled as a single unit to either be on or off.

While the present invention has been described with respect to a limited number of embodiments, those skilled in the art will appreciate numerous modifications and variations therefrom. It is intended that the appended claims cover all such modifications and variations as fall within the true spirit and scope of this present invention. 

1. An apparatus comprising: a first stage including: a plurality of master radio frequency (RF) devices each to receive a common incoming RF signal and provide an RF current, each of the master RF devices comprising at least one transconductor; a plurality of master local oscillator (LO) devices each coupled to an output of one of the master RF devices, each of the master LO devices to receive the RF current and mix the RF current with a master clock signal to obtain a mixed signal; a plurality of rotating switch devices each coupled to one of the master LO devices to cyclically switch the corresponding mixed signal to each of a plurality of mixer loads; the plurality of mixer loads each coupled to the plurality of rotating switch devices, each of the plurality of mixer loads to perform gaining and filtering of the corresponding mixed signal; a second stage including: a plurality of gain stages each coupled to at least one of the plurality of mixer loads to weight the output of the corresponding mixer load and to provide an output to a summer.
 2. The apparatus of claim 1, wherein each of the master RF devices has a programmable weight.
 3. The apparatus of claim 2, wherein the programmable weight is based on a number of master RF devices to be activated.
 4. The apparatus of claim 2, wherein each of the master RF devices is sine weighted.
 5. The apparatus of claim 2, further comprising N master RF devices and M gain stages, wherein N is less than M.
 6. The apparatus of claim 2, wherein each of the gain stages has a programmable weight proportional to a weighting of a corresponding one of the master RF devices.
 7. The apparatus of claim 1, wherein the number of the plurality of master RF devices and the number of the plurality of master LO devices is based on a frequency of a desired channel present in the common incoming RF signal.
 8. The apparatus of claim 7, wherein for a given value of N, at least one of the plurality of mixer loads is disabled.
 9. The apparatus of claim 1, wherein a first N/2 master RF devices are to receive a first differential polarity of the common incoming RF signal and a second N/2 master RF devices are to receive a second differential polarity of the common incoming RF signal.
 10. The apparatus of claim 1, further comprising an antenna to provide the common incoming RF signal to the plurality of master RF devices, wherein the antenna is to be coupled to the plurality of master RF devices without a low noise amplifier (LNA) or a tracking filter interspersed therebetween.
 11. The apparatus of claim 1, wherein at a first period of the master clock signal, the mixed signal from a first master LO device is to be cyclically switched to a first mixer load and the mixed signal from a second master LO device is to be cyclically switched to a second mixer load.
 12. The apparatus of claim 1, further comprising a counter circuit to generate a first pulse signal and a shift register coupled to receive the first pulse signal and the master clock signal and to output one of N rotational pulse signals to each of the plurality of rotating switch devices.
 13. The apparatus of claim 12, wherein each of the plurality of rotating switch devices comprises a plurality of rotating switches each comprising a transistor to receive the mixed signal from a corresponding one of the master LO devices and to output the mixed signal when the transistor is enabled by one of the N rotational pulse signals, wherein a value of the rotational pulse signals is changed when a value of the master clock signal is at a low state.
 14. An apparatus comprising: a multi-stage harmonic rejection mixer to receive and downconvert an incoming radio frequency (RF) signal to a second frequency signal, comprising: a first stage including: first and second sets of paths each to receive and process the incoming RF signal of a respective first and second polarity, each of the first and second sets of paths including: weighted transconductors having a weighting different than at least one other of the weighted transconductors to receive the incoming RF signal polarity and provide an RF current, a master local oscillator (LO) device coupled to an output of the transconductor to receive the RF current and mix the RF current with a master clock signal to obtain a mixed signal, a rotating switch device coupled to the master LO device to cyclically switch the corresponding mixed signal to each of a plurality of mixer loads coupled to the plurality of rotating switch devices of the first and second set of paths to perform gaining and filtering of the corresponding mixed signal; and a second stage having a third set of paths each including a gain stage coupled to at least one of the mixer loads to weight the output of the corresponding mixer load and to provide an output to a summer, the gain stage weighted proportionately to a corresponding transconductor of the first and second sets of paths, and the summer coupled to the plurality of gain stages to sum the weighted outputs to obtain the second frequency signal.
 15. The apparatus of claim 14, wherein a product of the transconductor weighting and the gain stage weighting is to reduce a gain error of the multi-stage harmonic rejection mixer.
 16. The apparatus of claim 14, wherein each of the plurality of rotating switch devices comprises a plurality of rotating switches each comprising a transistor to receive the mixed signal from a corresponding one of the master LO devices and to output the mixed signal, wherein each transistor is to be switched when a value of the master clock signal is at a low state to reduce a phase error of the multi-stage harmonic rejection mixer.
 17. The apparatus of claim 15, further comprising N weighted transconductors and M gain stages, wherein N is less than M.
 18. A method comprising: passing an incoming radio frequency (RF) signal through a plurality of weighted transconductors to generate a plurality of weighted RF signals, the plurality of weighted transconductors each having a weighting according to a first sine function; mixing the weighted RF signals with a master clock signal in a plurality of mixer devices to generate a plurality of weighted second frequency signals; cyclically rotating each of the plurality of weighted second frequency signals via a multi-phase bus to a plurality of mixer loads; outputting the plurality of weighted second frequency signals from each of the plurality of mixer loads to a corresponding one of a plurality of gain stages; generating a double weighted second frequency signal in each of the plurality of gain stages, wherein each of the gain stages is weighted to according to a second sine function; and combining the double weighted second frequency signals to output a second frequency signal.
 19. The method of claim 18, further comprising passing the incoming RF signals to N of M of the plurality of weighted transconductors based on a frequency of a desired channel within the incoming RF signal.
 20. The method of claim 18, further comprising reducing a gain error by performing the double weighting. 